1. Field of the Invention
This invention relates to an improvement in a spread spectrum receiver.
2. Description of the Prior Art
In recent years, those skilled in the art have focused on code division multiple access (CDMA) techniques using direct sequence/spread spectrum (DS/SS)signaling in a mobile communication field. To apply DS/SS communication to mobile communication, it is difficult to recover carrier, thus it is desired to make a quasi-coherent detection at a receiver.
For the quasi-coherent detection, if a local carrier has a frequency offset, the error rate characteristic degrades. Thus, an AFC (automatic frequency control) circuit is indispensable for frequency control of the local carrier or compensation of the effect of the frequency offset.
Referring to FIG. 1, the prior art is described. FIG. 1 is a block diagram showing the configuration of a conventional spread spectrum receiver. In FIG. 1, a received SS signal is quasi-coherently detected by a quasi-coherent detector and AFC circuit 100, and output as a complex baseband signal. This complex baseband signal is input to a correlator 110 which then calculates the correlation between the signal and a PN signal used for spread spectrum modulation of the received SS signal to generate a complex correlation signal. This complex correlation signal is input to an square absolute value circuit 120 which then outputs a synchronization setting signal having the square of the absolute value of the complex correlation signal. An initial acquisition and synchronization tracking circuit 130 uses the synchronization setting signal to generate a symbol clock synchronized with the period of the PN signal contained in the received SS signal and a chip clock synchronized with the chip duration of the PN signal.
On the other hand, the complex correlation signal output from the correlator 110 is also input to a demodulation processing circuit 140 which then performs demodulation processing of the signal conforming to the primary demodulation to generate demodulated data.
Next, the configuration and operation of the quasi-coherent detector and AFC circuit 100 are described. FIG. 2 is a block diagram showing the configuration of the conventional quasi-coherent detector and AFC circuit 100. In FIG. 2, a received SS signal is multiplied by a local carrier output from a voltage-controlled oscillator (VCO) 230 by a multiplier 210 and from the result, an image frequency component is removed through a low pass filter 250. Then the resultant signal is converted into digital data by an analog-to-digital (A/D) converter 270 to generate a real component of a complex baseband signal. Likewise, the received SS signal is also multiplied by a local carrier whose phase is shifted .pi./2 by a phase shifter 240 by a multiplier 240 and the result is passed through a low pass filter 260 and an analog-to-digital (A/D) converter 280 to generate an imaginary component of the complex baseband signal.
The complex baseband signal thus provided becomes an output signal of the quasi-coherent detector and AFC circuit 100 and is also input to an error signal generation circuit 290.
Next, the configuration and operation of the error signal generating circuit 290 are described. FIG. 3 is a block diagram showing the configuration of the conventional error signal generation circuit 290. A process in which the error signal generation circuit generates an error signal is described. Assume that the primary modulation is BPSK (binary phase-shift keying), that the period of a PN signal used for spread spectrum modulation is M chips, that the chip duration is T.sub.c, and that the value of the mth (m=1, . . . , M) PN signal is u.sub.m .epsilon.{-1, 1}. Also, assume that the symbol duration of data is T.sub.d =MT.sub.c, that the value of transmit data at time nT.sub.d (n is an integer) is a.sub.n .epsilon.{-1, 1}, and that the angular frequency of a transmit carrier is .omega..sub.C.
The receiver receives a SS signal having a value of a.sub.n u.sub.m cos [.omega..sub.C (nT.sub.d +mT.sub.c)] at time nT.sub.d +mT.sub.c.
Assume that the angular frequency of a local carrier used for quasi-coherent detection is .omega..sub.C +.DELTA..omega. and its initial phase is .phi.. Assume that the sampling period of A/D conversion equals the chip duration and that no quantization error exists. The value of complex baseband signal at time nT.sub.d +mT.sub.c =(nM+m) T.sub.c, r.sub.nM+m, is given by EQU r.sub.nM+m =a.sub.n u.sub.m exp [-j {.DELTA..omega.(nM+m) T.sub.c +.phi.}](1--1)
At the error signal generation circuit 290 shown in FIG. 3, a deviation signal generation circuit 400 outputs a signal of exp[j.omega..sub.0 t]. The signal is converted by a conjugate circuit 410 into a signal of exp[-j.omega..sub.0 t] which is a complex conjugate.
The complex baseband signal input to the error signal generation circuit 290 is multiplied by the signal exp[-j.omega..sub.0 t] by a multiplier 420 and positive angular frequency deviation .omega..sub.0 (.omega..sub.0 &gt;0) is given, then the result is output as a "positive deviation baseband signal." The complex baseband signal is also multiplied by the signal exp[j.omega..sub.0 t ] output from the deviation signal generation circuit 400 by a multiplier 430 and negative angularfrequency deviation -.omega..sub.0 is given, then the result is output as a "negative deviation baseband signal."
Assuming that the values of the positive and negative deviation baseband signals at time (nM+m) T.sub.c are r .sub.pnM+m and r .sub.pnM+m respectively, the values of r.sub.pnM+m and r.sub.nnM+m are given by EQU r.sub.nnM+m =a.sub.n u.sub.m exp [-j {(.DELTA..omega.+.omega..sub.0) T.sub.c +.phi.}] EQU r.sub.nnM+m =a.sub.n u.sub.m exp [-j {(.DELTA..omega.-.omega..sub.0) T.sub.c +.phi.}] (1-2)
The positive and negative baseband signals are input to complex correlators 440 and 450, respectively, which then calculate the correlations between the signals and PN signals to generate a "positive deviation correlation signal" and a "negative deviation correlation signal." Assuming that the values of the positive and negative deviation correlation signals corresponding to transmit data a.sub.n provided every symbol duration T.sub.d are c.sub.pn and c.sub.nn respectively, from Equation (1-2), c.sub.pn and c.sub.nn are given by ##EQU1##
Further, square absolute value circuits 460 and 470 square the absolute values of the positive and negative correlation signals to generate a "positive deviation error signal" and a "negative deviation error signal." Last, the negative deviation error signal is subtracted from the positive deviation error signal by a subtractor 480 and the resultant signal is latched by a latch 490 every symbol duration T.sub.d, thereby providing an error signal. That is, the error signal e.sub.n corresponding to the transmit data a.sub.n is given by ##EQU2##
By setting the value of given angular frequency deviation to the range of 0&lt;.omega..sub.0 .ltoreq.2.pi./T.sub.d, the error signal e.sub.n has value corresponding to angular frequency offset .DELTA..omega.. FIG. 4 shows the relationship between the error signal e.sub.n and phase rotation amount .DELTA..omega.T.sub.d for the symbol duration when M =127 and .omega..sub.0 =.pi./T.sub.d. From the graph, it is understood that the error signal e.sub.n is substantially proportional to the angular frequency offset .DELTA..omega. in the range of .vertline..DELTA..omega.T.sub.d .vertline..ltoreq..pi.. Thus, the error signal corresponding to the frequency offset can be provided by the error signal generation circuit 290 in FIG. 3.
Referring again to FIG. 2, the configuration and operation of the quasi-coherent detector and AFC circuit 100 are described. The error signal provided by the error signal generation circuit 290 as described above, e.sub.n, is multiplied by gain .alpha. by a multiplier 300, then the result is integrated by an integrator 310 to improve the signal-to-noise ratio of the error signal. The output signal of the integrator 310 is converted into an analog signal by a digital-to-analog (D/A) converter 320, and the resultant voltage signal is used to control the VCO 230 which oscillates local carrier, thereby performing AFC operation so as to always set the angular frequency offset .DELTA..omega. to 0.
Since the conventional spread spectrum receiver is configured as described above, in addition to the correlators used for PN signal synchronization and data demodulation, another correlator is required for the AFC error signal generation circuit. Thus, the configuration is prone to become complicated and miniaturization and low power consumption are not easy to accomplish.